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Shipilov direct conversion receiver. Three-band direct conversion shortwave receiver. For the scheme "Experimental detector VHF-microwave receivers"

Direct conversion receivers (DCC), more precisely heterodyne receivers, began to be used by radio amateurs relatively recently - from the late 60s - early 70s of the last century. They very quickly gained wide popularity due to the simplicity of the circuit and the high quality of work. Simple (on several transistors or one or two microcircuits) one-two-band designs of two-band PPPs, accessible for repetition even to novice radio amateurs, were especially popular. As a rule, having high sensitivity, these receivers had a relatively small dynamic range for crosstalk (DD2) - the AM suppression coefficient, with rare exceptions, did not exceed 70-80dB. Attempts to increase DD2 and suppress the second band by at least 30-40 dB led to such a complication of the design that mass repetition was out of the question.

Thanks to the appearance on the wide market of new high-speed digital microcircuits and high-quality low-noise op-amps, it became possible to implement a new approach to building single-sideband PPPs, using digital switches as a mixer and using well-established circuitry of functional units on the op-amp in the rest of the circuit. This approach makes it possible to ensure good repeatability, guaranteed high PPP parameters and, at the same time, to abandon such low-tech elements as multi-turn inductors, balancing transformers and almost completely eliminate tuning elements and labor-intensive adjustment work (of course, with the exception of tuning the PDF and GPA circuits). The price for this is an increased number of microcircuits and the need for preliminary selection (if there are no appropriate precision ones) of some resistors and capacitors, which, however, is easy to do using an ordinary Chinese digital camera.

An experimental sample of a single-band PPP, brought to your attention, is an illustration of one of the possible variants of a circuit design based on a modern element base.

main parameters
Operating frequency ranges, MHz — 1.8, 3.5, 7

Receive path bandwidth
(in terms of level - 6dB), Hz - 400-2900

Sensitivity of the receiving path from the mixer input
(bandwidth 2.5 kHz, S / N ratio - 10 dB), μV, not worse - 0.7 *

Cross-modulation dynamic range (DD2) at 30% AM and detuning 50 kHz, not less than, dB — 110*

Adjacent channel selectivity
(when detuned from the carrier frequency by -5.9 kHz + 3.7 kHz), not less than, dB - 60

Suppression of the upper sideband, not less than, dB - 41

Squareness factor through frequency response

(by levels -6, -60dB) - 2.2

AGC adjustment range when the output voltage changes by 12 dB, not less than, dB - 72 (4000 times)

Output power of the low-frequency path at a load of 8 ohms, less than, W 0.8

The current consumed from the external stabilized

power supply 13.8V, no more, A - 0.4

* this figure is limited by the capabilities of the equipment used for measurements and, in reality, can be higher.

Node A2 is a local oscillator based on a single, non-switchable oscillator at frequencies of 28-32 MHz with electronic frequency tuning by a multi-turn resistor and a frequency divider with a variable division factor of 1,2,4. The necessary stability with the help of the CAFC and the digital frequency reading is provided by the A5 node, made on the basis of the ready-made Makeevskaya digital scale, which can be purchased in many regions of Ukraine and Russia and is not described here, as an option for self-production, one can recommend the well-proven development A. Denisov [5]. The main processing of the signal - its transformation, suppression of the upper sideband and filtering is performed by node A3. To obtain good selectivity, the principle of successive selection is applied, when, in addition to the main active band-pass filter, in fact, in each amplifying stage, the bandwidth is limited at the level of 300-3000 Hz by the appropriate choice of the denominations of isolation capacitors and in the OOS circuits.

To suppress the upper sideband, the method described in detail in and based on the use of a 6-section phase shifter in a 4-phase signal system is used, which allows relatively simple means, despite the increased number of elements, to obtain good suppression and high temperature and time stability of the parameters. For getting

A 4-phase signal system uses a digital phase shifter, which greatly simplifies the creation of multi-range structures.

The signal from the PDF output goes to the mixer, which is an inexpensive and affordable eight-channel switch 74NS4051 with an average switching time of 20-22nS. The motivating reason for this choice was the phenomenal values ​​of DD obtained by radio amateurs when testing microcircuits 74NS4066, 74NS4053 of the same series as mixers. Experiments carried out during the development of this receiver confirmed the high dynamic parameters of the mixer based on 74HC4051. According to my estimates, the potential DD2 (the level of AM suppression - namely, it determines the dynamic range of acceptable signals for the PP) for 74NS4051 at frequencies up to 7-8 MHz is about 134-140 dB, is limited from above by AM interference levels of 300-400 mV, and from below by the internal noise of the switch , which are less than 0.05 μV.

In the experimental receiver offered to the attention of readers, the DD2 level of 110dB is limited not by the mixer, but by the preliminary ULF, from above due to the direct detection of AM interference in the preliminary ULF, and can be improved by 10-20dB by installing additional low-pass filters after the mixer, but from the bottom by the noise of the preliminary ULF, implemented, like all other nodes, on an inexpensive and affordable dual low-noise (noise spectral density less than 5nV / Hz) op-amp NE5532. The use of less noisy op-amps, for example, LT1028 with 1nV / Hz, will improve the sensitivity by a factor of 3-4; increase DD2 by another 10-12dB.

The use of an eight-channel switch as a mixer (in our case, only half - four channels) 74NS4051 made it possible to simplify the circuit, due to the fact that the functions of the phase shifter are performed by the internal control logic of the switch, the address inputs of which receive control signals from the counter to 4. In this case, the local oscillator frequency should be four times higher than the operating frequency. As a result, a 4-phase signal system is formed at the mixer output, which, after preliminary amplification, is fed to a 6-link phase shifter. Further, the signal of the lower sideband, which received a zero phase shift, is summed at the adder, and the mirror upper band, which received a phase shift of 180 degrees, is subtracted and suppressed. The main active band-pass filter is connected to the output of the adder, which is a successor of the included high-pass filter of the 3rd and low-pass filter of the 6th order.

The filtered useful signal is fed to node A4, which consists of a voltage-controlled amplifier, an intermediate amplifier and a final ULF, to the output of which a loudspeaker is connected, an AGC detector and gain and volume controls.

A schematic diagram of node A3 - the main block for receiving and processing a signal is shown in Fig. 2. Further in the text, positional designations of parts of functional units A2, A3, A4 (Fig. 2-4) will have additional indexing (2C1, 3C1, etc., respectively), which is conventionally not shown in these figures. Positional designations of hinged parts on the diagram of interconnections of the receiver fig. 5 are not repeated, so references to them are given without additional indices.

The signal from the output of the range filter (not shown in the diagram, as already noted, in this capacity the author used the preselector described in) through the matching transformer 3Tr1 goes to the 3R5 resistor and then to the 3DD1 4-phase mixer, made on the basis of the 74NS4051 eight-channel switch. To increase the speed of the switch, the 3DD1, 3DD2 microcircuits are powered by an increased supply voltage of + 8V from the 3DA5 stabilizer, which seems to be quite acceptable, because. experience shows that microcircuits of the 74HC, 74AC series work reliably when the supply voltage is increased to 10V.

Resistor 3R5 improves the balance and equalizes the resistance of the open state of the keys, which have a resistance of about 50 ohms with a technological spread of +-5 ohms. The bias voltage is applied to the switch input through the 3R6 resistor, which is formed at the midpoint of the 3R3 3R4 resistive divider and is equal to half the supply voltage, which provides its operation in the most linear section. The control signals to the switch come from a synchronous counter-divider by 4, made on the D-flip-flops of the 3DD2 74NS74 microcircuit, connected according to the Johnson ring circuit. Despite the external similarity with the digital phase shifter proposed by V.T. Polyakov, in this circuit its main function is a counter.

The functions of the phase shifter are performed by the internal control circuit of the switch itself, since applied non-standard inclusion, for clarity in Fig. 2 opposite the corresponding pins of the 3DD1 chip, the phases of the output signal are indicated. Load capacitors are connected to the output of each of the 4 phase channels, effectively isolating the useful signal and suppressing the conversion by-products. The reason for this efficiency is that this 4-phase switch mixer + capacitors is an example of a classic digital filter (or switched capacitor filter if you prefer). Taylor was the first to describe and patent this circuit solution in relation to mixers, and this circuit is called the Taylor detector.

Where Rist, Ohm, is the sum of the resistances of the antenna circuit 50 ohms, transformed by 3Tr1 9 times, i.e. 450 ohms, the resistance of the public key (about 50 ohms) and the resistor 3R5, Snagr is equal to the sum of the capacitors 3C8,3C9 in farads, and n \u003d 4 is the number switched capacitors. In our case, the calculated value of the cutoff frequency of 3400 Hz, on the one hand, provides good suppression of out-of-band interference, and on the other hand, introduces a noticeable additional phase shift into the useful signal, so the corresponding capacitances in all 4 channels must be thermally stable and selected with an accuracy of no worse than 0.5 % (hereinafter, the accuracy of selecting elements of 4 channels among themselves is implied, the absolute value can have a spread of up to 5%). These requirements are met by low-frequency capacitors of the MBM, K71, K73 series, etc., and for effective HF filtering, they are connected in parallel with relatively small ceramic capacitors (possible values ​​​​1000-4700pF) with thermal stability no worse than M1500.

To the load capacitors of the mixer through separating capacitors 3C10, 3C13, 3C16, 3C19 of large capacity (at first glance, the use of separating capacitors after the mixer is unnecessary, because in a perfectly working mixer, the voltage across the load capacitors is the same, but in practice due to some asymmetry channels, a small noise voltage appears, which increases the total noise by 2-3 times when preamplifiers are directly connected), which must be non-electrolytic, preamplifiers 3DA1, 3DA2 are connected, connected according to the differential measuring amplifier circuit, further improving the symmetry of the signals and suppressing common-mode interference ( AM detection products, interference with mains frequency, etc.) in proportion to Kus=1+2*(3R12/3R11), in this case, 13 times. This value of pre-amplification is optimal in the opinion of the author in order to compensate for losses in a 6-link phase shifter. Resistors in feedback circuits 3R11….16 must be selected with an accuracy of at least 0.5%. A 4-phase 6-link RC phase shifter on elements R17-R40 and C21-C44 is connected to the outputs of the differential preamplifier. Such a phase shifter, despite the increased number of elements, is simple in design. Due to the mutual compensation of phase and amplitude imbalances of individual chains, it is possible to use elements with a tolerance of + -5% of the absolute value (of course, the selection accuracy in quadruples should be no worse than 0.5%) while maintaining high phase shift accuracy. With the values ​​of the elements indicated on the diagram, the calculated value of the suppression of the mirror sideband in the frequency range of 300-3300 Hz is about 50 dB, but practically due to the spread in the values ​​of the elements and the final impedance of the adder, the suppression is 41-43 dB. Further, a 4-phase signal is fed to the inputs of the 3DA3.1 adder, made on the basis of a differential amplifier with an input resistance of 330 kOhm and a gain of 10,

where, due to the obtained phase shifts, the signals of the lower sideband are added and amplified, and those of the lower sideband are subtracted and suppressed. An active main filter of the signal frequency is connected to the output of the adder, which is made on three successively connected links of the 3rd order - one high-pass filter with a cutoff frequency of 350 Hz on the op-amp 3DA3.2 and two low-pass filters with a cutoff frequency of 3000 Hz on the op-amp 3DA4.1 and 3DA4.2, respectively.

To improve isolation and reduce noise in the power supply circuit, the adder and filter stages are powered through a separate 3DA6 integrated regulator. The 3R52,3R57 supply voltage divider provides a bias voltage for normal operation of the op-amp 3DA3.2, 3DA4 with a unipolar supply.

The filtered signal from the output X9 of node A3 is fed to the input X1 of node A4, the circuit diagram of which is shown in Fig. 3, and through the isolation capacitor 4C2 to the adjustable amplifying stage on the op-amp 4DA1.1. Its Kus is determined by the ratio of the total resistance of the 4R4 resistor connected in parallel in the OOS circuit and the resistance of the drain-source channel of the 4VT1 KP307G field-effect transistor (here you can use any transistors from the KP302,303,307 series that have a cut-off voltage of no more than 3.5V at the maximum initial drain current) to the 4R2 resistor and when the bias voltage at the 4VT1 gate changes from 0 to + 4V, it changes in the range from 3 to 0.0005 times or + 10 ... -66 dB, which allows you to apply effective automatic (AGC) and manual adjustment of the total receiver gain (a kind analog of adjustment for HF, IF in superheterodynes). The 4R5,4R7,4C4 chain supplies half the signal voltage to the 4VT1 gate, which improves the linearity of the field effect transistor control characteristic, as a result of which, even with an input signal of 2eff (the maximum possible signal at the output of the main bandpass filter), the level of non-linear distortion does not exceed 1%.

The signal from the 4DA1.2 output, which provides a gain of 50 for normal operation of the AGC, enters through a passive band-pass filter 4С13.4R12.4C15, which reduces the excess gain by 4 times to the volume control R and then through a single-link low-pass filter (4R16.4C17) to the input of the final ULF 4DA3 LM386 with Kus=20.

The signal from the 4DA1.2 output through the 4C12,4R11 chain goes to the AGC detector, which is made on 4VD1-4VD5 diodes and has two control circuits - an inertial one on the 4C8 capacitor and a relatively fast one on the 4C9 capacitor, which makes it possible to improve the operation of the AGC in conditions of impulse noise. The common connection point of the AGC detector elements is connected to the supply voltage divider 4R13, 4R14, which creates the initial bias voltage of the field-effect transistor. The 4R15 trimmer resistor sets the optimal initial bias voltage for a particular transistor instance and, if necessary, corrects the initial value of the total receiver gain. Resistor Rrf carry out operational adjustment of the overall gain.

To improve isolation and reduce noise in the power circuit, the input stages are powered through a separate integrated 4DA2 stabilizer. The 4R1,4R3 supply voltage divider provides a bias voltage for normal operation of the 4DA1 op-amp with a unipolar supply.

Schematic diagram of node 2 (GPA) is shown in fig. 4

A slightly modernized GPA circuit from the YES-98M transceiver based on the Kolpitz generator was taken as the basis. The active element of the GPA - the 2VT2 transistor is connected according to the emitter follower circuit, due to the high input resistance and the small capacitance of the 2C11 capacitor, the shunting of the oscillatory circuit is insignificant. The generator, assembled according to the Kolpitz circuit, is known for its stable generation, and two branches of negative feedback: parallel (resistor 2R12) and serial (resistor 2R14) ensure the operation of the 2VT2 transistor in the mode of a constant (thermostable) current generator. The low capacitance of the emitter junction of the KT368A transistor (about 2 pF) and the low output impedance of the cascade create conditions for a good decoupling of the oscillatory system as a whole from the subsequent load. The collector capacitance 2VT2 (about 1.5 pF) is many times smaller than the 2C8 capacitor, and has no effect to the oscillatory system. The use of a low-noise transistor KT368A (with a normalized noise figure) and the above features contributes to the creation of a generator with good thermal stability and a low level of side (phase) noise. provides good decoupling of the master oscillator from subsequent stages.

Elements 2DD1.1 and 2DD1.2 form a rectangular signal. Triggers 2DD2.1 and 2DD2.2 are designed to divide the GPA frequency by 2 or 4 for the ranges of 3.5 or 1.8 MHz, respectively. The encoder, assembled on diodes 2VD7 ... 2VD9 and elements of microcircuits DD1 and DD3, when applying a range voltage of + 13.8V, ensures the selection of the corresponding subrange. In this case, the triggers not participating in the division are blocked, which eliminates the appearance of interference from them at the receive frequency. From the output DD3.3, the signal is fed to the counter of the converter unit (input X3 of node A3). Frequency tuning is carried out by KV132A varicaps and a multi-turn potentiometer SP5-39B, although the disadvantages of this tuning method are well known. The traditional tuning method with a variable capacitor is, of course, preferable, and its quality indicators are higher.

The 2R1, 2C2 2R5, VD3, 2C5 chain is part of a digital automatic frequency control (DAC) circuit implemented using the Makeevskaya digital scale, which allows you to work not only SSB and CW, but also digital modes of communication

The generator itself operates in the frequency range from 28 to 32 MHz.

It should be noted that on the 40-meter range, the tuning interval of the receiver is too wide and is 1 MHz, which leads to a high tuning density, therefore, for using a tuning resistor 2R4, it is limited to 28.0 ... 28.8 MHz (7-7.2 MHz). On the ranges of 1.8 and 3.5 MHz, this resistor is shunted with an open key on the transistor 2VT1 (it is possible to use KT208, KT209, KT502 with any letter index), which closes when a control voltage of +13.8V is applied from the range switch to the 7 MHz output Transistor 2VT2 is selected for maximum gain, not less than 100. For the selection of loop capacitors, capacitors with different TKEs are required: MPO, P33 and M47. As 2DD1, 2DD3, you can use the TTL series 555LA4, and instead of

2DD2 - 555TM2, high-speed CMOS KR1554LA4, KR1554TM2, or 74NS10 and 74NS74, respectively. Diodes KD522 can be replaced by almost any silicon high-frequency diodes with low reverse currents (for example, KD503, KD521).

Figure 5 shows the interconnection diagram of the receiver. All board-to-board connections of high-frequency circuits are made with a thin coaxial cable, and low-frequency circuits with a conventional shielded one. The voltage regulator of the digital scale DA1 (Kren 5A or 7805) heats up slightly (the current consumption with imported ALS is not more than 200mA.), So you can screw it in any convenient place in the case. Quenching resistor R2 with a power of at least 2W. Variable resistors R1 (Setting), R3 (Volume control), R4 (Gain control) and switches SA1 (Enable Attenuator -20dB), SA2 (range switch), SA3 (Enable DAC) are located on the front panel. The boards in the receiver case are mounted on metal racks, but this does not exclude an additional "ground" bus that connects all the boards to each other.

About the details. As noted above, for successful repetition, some positions of the resistors and capacitors of block A3 require preliminary selection. With the help of a digital ohmmeter, for example, the Chinese digital camera, it is easy to match pairs or fours with an accuracy of up to the third decimal place, taking into account the fact that, as a rule, the absolute value can have a spread of up to 5%. Many models of multimeters also have capacitance measurement modes, which will make it easy to select capacitors. For the selection of capacitors, the author used an attachment to the frequency meter to measure the inductance, connecting to it an inductance coil with a few tens of μH. After that, connecting "on weight" capacitors, we select those that give close frequency values. The spread of values ​​​​for capacitors from one factory batch is small. If the capacitors are from one box, then, as a rule, out of a dozen it was possible to pick up two fours with an accuracy of no worse than 1%. Despite the apparent complexity of the selection, the author spent no more than an hour on the selection of all four resistors with an accuracy of 3 digits and capacitors with an accuracy of 2 digits.

The phase shifter capacitors must be thermally stable, in no case should low-frequency ceramics of the TKE H30, H70 and H90 groups be used (the capacitance of the latter can change almost 3 times with temperature fluctuations). You can use metal-paper MBM, film and metal-film series K7X-XX. It is desirable to use the same types of capacitors as part of active filters and separation filters in ULF cascades, because. they determine the frequency response. In this case, the allowable spread of ratings can be 10%, and in these nodes, with great success, you can use instances that have not passed the selection for the phase shifter.

Blocking ceramic and electrolytic can be of any type.

Coil L1 with an inductance of about 0.8 μH of a smooth range generator is wound on a ribbed ceramic frame with a diameter of 12 mm. It has 12 turns of PEV-2 wire 0.5-0.7 mm, laid in a groove with a 1 mm pitch and placed in a screen, which can be used, for example, as a housing from a RES-6 relay.

The 3Tr1 matching transformer contains 15-18 turns of a wire folded three times with a diameter of PELSHO (you can also use PEV, PEL) 0.1-0.25 mm with a small twist (3 twists per cm) on a ferrite ring with a diameter of 7-10 mm with a permeability of 1000-2000 High-frequency chokes - DM-0.1 with a nominal value of 50-200 µg, they can be wound on ferrite rings with a diameter of 7-10 mm with a permeability of 1000-2000, 25-30 turns are enough with a wire with a diameter of 0.15-0.3 mm.

Parts installed by the method of hinged mounting on the chassis (see Fig. 5) can be of any type. The exception is the multi-turn variable resistor R1 SP5-39B. This resistor must be of high quality. The instability of the resistance, the unevenness of its change will significantly impair the performance of the receiver. If necessary, it can be replaced by two conventional potentiometers connected according to Fig.6.

Special requirements for the rest of the details, if any, are stated above when describing the nodes.

Construction and installation. Most of the receiver parts are mounted on three printed circuit boards, corresponding to its three blocks A2 (Fig. 7), A3 (Fig. 8), A4 (Fig. 9), made of double-sided foil fiberglass. The second side serves as a common wire and screen. Holes around the leads of parts that are not connected to a common wire should be countersinked with a drill with a diameter of 2.5-3.5 mm. The conclusions of the parts connected to the common wire are marked with a cross. An archive with author's drawings of printed circuit boards in lay format can be

Photos of mounted nodes and the receiver as a whole






Setting up the receiver should start from node A2 of the GPA, which is disconnected from the main node for the period of adjustment. First, you need to apply a voltage of about 2.7V to the 2X1 output from the auxiliary divider and short-circuit the 2C12 capacitor with a jumper. Having applied the supply voltage, by selecting the 2R12 resistor, set the voltage at the emitter of the 2VT2 transistor to about 1.4-1.6V when used as 2DD1 TTL series 1533LA4.555LA4 or 2.3-2.6V if CMOS KR1554LA4.74NS10 are used. After that, you can remove the jumper and apply a control supply voltage to the 2X8 output (enabling the 1.8 MHz range). A digital scale or frequency meter is connected to the output of the GPA (output 2X12) through a resistor with a resistance of 200 ... 300 Ohms. By moving the slider of the resistor R1 to the upper position according to the diagram, by selecting the capacitor 2C12 and adjusting the 2C10, the generation frequency is set slightly below 7000 kHz (by 5 ... 10 kHz). Then the engine of the resistor R8 is transferred to the lower position according to the scheme. The operating frequency should be slightly above 8000 kHz. If this cannot be done and the overlap is less, then a larger capacitor 2C9 should be installed and vice versa, if the overlap is greater, the capacitance of the 2C9 capacitor should be slightly reduced. Since the capacitance of this capacitor somewhat affects the frequency of the GPA, after changing its value, the frequency overlap of the GPA should be checked again. Having achieved the required value in the 1.8 MHz range, the GPA is transferred to the 7 MHz range by applying a control supply voltage to the 2X9 output. Then the engine of the resistor R8 is moved to the lower position according to the scheme and by adjusting the resistor 2R4 the generation frequency is set slightly above 28800 kHz. In the author's version, loop condensers with TKE M47 were used and no additional thermal compensation was performed. At the same time, at 7 MHz, the initial frequency overshoot for the first 2 minutes did not exceed 800 Hz, and then the frequency instability was less than 100 Hz for 15 minutes. When the DAC was turned on, the frequency remained unchanged for several hours.

The main signal processing unit (node ​​A3) and ULF (node ​​4), when using parts of the required ratings and the absence of errors in the installation, do not require adjustment.

The last step in setting up the receiving path is setting the AGC threshold and gain control limits. To do this, the sliders of the resistor R3 Volume and the resistor R4 Gain (see Fig. 5) are set to the left position according to the diagram, and the slider of the 4R15 tuning resistor is set to the right.

Connect a 50 ohm resistor to the input of the receiver.

An oscilloscope or an avometer is connected in parallel to the speaker (pins 4X7,4X8) in the AC voltage measurement mode to the receiver output.

By moving the slider of the 4R15 tuning resistor, find the position at which the noise begins to decrease and by moving further set the noise level, which still “does not press on the ears” (according to the author, about 30-40mV). This will be the optimal setting for the AGC threshold (the start of operation is about 2-3 μV) and the total initial gain (about 120-150 thousand).

Bibliography

  1. Tietze W., Shenk K . Semiconductor circuitry. - M.: Mir, 1982,.
  2. Horowitz P., Hill W . The Art of Circuitry: Volume 1. - M.: Mir, 1983
  3. S. Belenetsky. Simple preselector for multi-band receiver . Radio, 2005, No. 9, pp. 70-73 or
  4. V. Abramov (UX5PS)C. Telezhnikov (RV3YF) Shortwave transceiver “Druzhba-M”. http://www.cqham.ru/druzba-m.htm .
  5. A. Denisov. The digital scale is a frequency meter with an LCD display and auto-tuning of the frequency. http://ra3rbe.qrz.ru/scalafc.htm
  6. Polyakov V . Radio amateurs about the direct conversion technique. - M.: Patriot, 1990.
  7. R.Green. “Bollet-proof” rf mixer.-“Electronics Word+Wireless Word”, No. 1/99, p.59

8. "Ideal" mixer for direct conversion receiver G. Bragin http://www.cqham.ru/trx41_01.htm

9.D.Tayloe, N7VE, “Letters to the Editor, Notes on “ideal” Commutating Mixers (Nov/dec 1999), “QEX, March/April 2001, p/61

  1. G.Bragin. Upgraded VPA for the transceiver "YES-98M. ― Radio Design N 14, p.3-7

11. Attachment for measuring inductance in the practice of a radio amateur. S. Belenetsky.-Radio, 2005, No. 5, p.26

j.Radio, 2005 №10, 11

Receiver development. As noted in the description of the receiver, due to the finite resistance of the adder, the degree of suppression of the mirror sideband is much lower than the theoretical one (this is especially noticeable in multi-link phase shifters-polyphasers). The main way to improve the performance of the polyphaser (up to theoretical limits) is to increase the input impedance of the adder by orders of magnitude (!) For example, by using voltage followers on the op-amp or on field devices. In the process of further tests and experiments with the receiver, the circuit was refined, allowing EASILY achieve suppression close to the theoretical limit. In this case, the circuit and design of the receiver is even slightly simplified.
To do this, you need (see the diagram in Fig. 2 or J. Radio, 2005, No. 10 p. 61-64) to remove resistors R41, R45 and capacitor C46, ​​increase resistor R46 to 33 kOhm, and replace resistor R44 with a wire jumper. On the printed circuit board (see Fig. 8), you should break the connection (cut the tracks) in 2 places

1.between the points connecting R37, C42 and R38, C43
2. between the points connecting R39, C44 and R40, R42, C41.
The signal is now taken from the phase shifter at one point through the non-inverting input of the op-amp (input resistance is at least a hundred MΩ). Wherein MEASURED coefficient transmission is close to 1. Interestingly, in this scheme, an additional adder is not needed, because single-sideband signal is good its quality already FORMED(!!!) in the phase shifter itself. Moreover, regardless of the signal pickup point, I tried to pick up the signal from all four chains, of course, in turn. For the first time, such a circuit design flashed on http://www.hanssummers.com/radio/polyphase/
And frankly, I did not pay serious attention to him -
the documentation is made by hand, in pieces - I thought that the author was too lazy to finish drawing 3 more op amps at the output of the phase shifter. While he himself was not convinced in practice - it works and works well!
Of course, in a certain sense, this is a compromise solution that allows one to obtain good results in the receiver by simple means at the cost of abandoning the classical method of signal pickup. At which (here I will allow myself to quote an explanatory comment by V.T. Polyakov from personal correspondence on how to remove signals from a polyphaser) “if you also remove the signal from the output of the PV that is opposite in phase, invert it and add it to the first one, then the output voltage will double. And moreover, if the remaining two outputs are connected to those already used, the output voltages will be less dependent on the load of the PV. Apparently, the creator of this FV with the completely unpronounceable surname Gschwindt in Russian, who published the scheme either in a German or in a Hungarian magazine in the 70s, reasoned like that.

After such refinement, the total Kus turns out to be about 130-150 thousand, the level of intrinsic noise at the output is approximately 27-30mV - the optimal values ​​\u200b\u200bin my opinion and do not need to be adjusted. you can download a variant of printed circuit board drawings from Pavel Semin ( syomin) performed in Sprint Layout 4.0 already taking into account this refinement, in which it was possible to slightly reduce the size of the boards.

Since the publication of the description of the receiver, several colleagues have already repeated the design and were satisfied with the quality of this receiver. Below, also as an example, are photos of the design by Igor Tredit ( Robin). Igor made a version of Pavel Semin's printed circuit board.

An important point - Igor, when repeating the receiver, encountered a small problem (this is the only case known to me, but I want to consider this issue in more detail - it may be useful to someone) - due to insufficient amplitude (less than 0.25V eff) at the GPA output when the range is turned on 7 MHz unstable, up to self-excitation in the microwave, triggers 74NS74 worked. The reason, in my opinion, was in the combination of an unsuccessful instance of 1533LA4, the gain of which drops sharply at frequencies of the order of 29-30 MHz and the bias voltage of the trigger DD2.1 (see Fig. 2), which, due to the spread of resistances R1, R2, may differ from optimal. The best way would be to put a more successful copy of the DD3 chip (see Fig. 4) or “play around” with the values ​​​​of R1, R2 (see Fig. 2), but this is easy to do if the microcircuits are installed on panels. But what if they are soldered to the board? It remains to select the offset by the values ​​of R1, R2, or do as Igor did. Leaving the switch supply voltage the same - 8V, he reduced the supply voltage of the DD2 chip to 6V, thereby increasing the relative amplitude of the GPA signal in relation to the trigger threshold, which is almost directly proportional to the trigger supply voltage.

The easiest way to do this is to supply power to DD2 through a 62-100 ohm resistor (selected according to the stable operation of triggers in the 7 MHz range). The last one you need to include in the gap of the printed conductor (see Fig. 8) between the foot 16 DD1 and capacitor C2.

Igor did not select capacitors for the polyphaser-phase shifter - he supplied them from one batch. Nevertheless, the degree of suppression of the upper side turned out to be high - which means that the design has a certain technological margin. Igor ( Robin) is very satisfied with the work of the receiver. When conducting a comparative listening to the air on Radio-76M2 and this PPP, he prefers the latter, noting its special softness of sound and the transparency of the air.

Finally I want to thank my colleagues and like-minded people on the forum http://forum.cqham.ru/viewtopic.php?t=4032

(Valery RW3DKB, Sergey US5QBR, Andrey WWW, Pavel syomin, Yuri UR5VEB, Alexander T, Oleg_Dm., Tadas, Alexander M, Alex007, Kestutis, US8IDZ, K2PAL, Victor, Igor Robin and many others) dedicated to the problems and development of T /PPP, those whose enthusiasm and downright fanatical love for the DIRECT TRANSFORMATION TECHNIQUE awakened in me, and in many others, the interest and desire to re-engage in PPP, those who carefully and tirelessly supported a real waterfall of information from around the world about new products and approaches , modern concepts, methods and circuit implementations of the PP technique. Thanks to all you friends. There are already many of us - fans of the DIRECT CONVERSION TECHNIQUE.

I can note with satisfaction that the design turned out to be really easy and affordable in repetition, while the parameters are excellent, not worse than the declared ones!

For example, a colleague Oleg Dmitrievich Potapenko, who has the possibility of instrumental measurements after a thorough measurement, received a sensitivity of 0.6 μV, DD2 of the order of 107-109 dB and suppression of the upper side - more than 54 dB). Of undoubted interest are his results of measurements of DD3 SPP by the two-frequency method, for which they used

generators with low phase noise IFR2040 from Aeroflex (aka IFR, even earlier it was Marconi).
1. We connect two GSS IFR2040 to the PPP through an adder with attenuation of 3 dB.
Both oscillator outputs are disabled - OFF
We measure the noise voltage at the PPP output with a V3-38B millivoltmeter.
Ush=19.5mV
2. Measure sensitivity
Setting up generators
F1 = 3.3329 MHz (working) output - ON (enabled)
F2=3.4349 MHz (interference2) output - OFF (disabled)
We give a signal Uc1 \u003d -111.8 dbm, at which Uout \u003d 62 mV (S/N \u003d 10 dB)
If we add 3 dB of the adder, we get

S=-114.8 dbm at S/N=10 dB.

3. We turn on interference with a spacing of 50 kHz, we receive at a frequency of 2F1-F2 \u003d 3.3329 MHz
F1=3.3839 MHz (interference1) output - ON
F2=3.4349 MHz (interference2) output - ON
Set equal signal amplitudes
Uс1=Uс2=-13.3 dbm, at which Uout=62 mV
4. Calculate DD3 = -13.3-(-111.8) = 98.5 dB

II. For 20 kHz spacing

F1=3.3539 MHz (interference1)
F2=3.3749 MHz (interference2)
Uс1= Uс2=-14.3 dbm and DD3 = -14.3-(-111.8) = 97.5 dB

After that, I carried out sensitivity measurements without an adder
1. We short-circuit the PPP input through 51 Ohm Ush = 17.5mV
S = -116 dbm, at S / N = 10 dB (Uout = 55 mV)
2. For a spacing of 50 kHz, I measured DD3 again
Uс1= Uс2=-14 dbm (or 44.6 mV) at which the output is 55 mV
DD3 = -14 -(-116) -3 = 99 dB

Receiver without housing, without shielding, self-made quartz local oscillator with a quartz double-crystal filter at the output, power supply B5-29 (+14 V). The signal was fed without a DFT, directly to the input trance of the mixer.
Obviously, precisely because of the lack of screening, the values ​​of Ush, S are somewhat floating from measurement to measurement.,

The dual-band direct conversion receiver is assembled on only two microcircuits and three transistors, but has good performance characteristics. Thanks to the use of a band-pass filter at the input (instead of a single circuit), good selectivity is achieved for the mirror and side reception channels.

The input stage on the field-effect transistor VT1 allows you to get high sensitivity (at least 0.5 μV) and, in addition, does not load the L3-C4 bandpass filter circuit and allows you to get excellent matching with the UHF input of the DA1 microcircuit.

In the microcircuit, in addition to RF amplification, the received signal and the signal of the smooth range generator are mixed. As a result of the conversion, an audio frequency signal is emitted on the primary winding of the transformer T1. The transformer (matching, from any pocket receiver) plays the role of a low-pass filter, the cutoff frequency of which is 2.5-3 kHz and is set by selecting the capacitance of the capacitor C20.

From the secondary winding, the signal is fed to the input of the low-frequency amplifier chip DA2, which has a large gain. It is reliable, does not get excited and does not overheat. Amplifier load can be an 8 ohm driver or headphones. The volume level is set using a variable resistor R14.

From the ULF output, through the resistor R12 and the rectifier on the diodes VD4 and VD5, the AGC voltage is applied to pin 9 of the DA1 microcircuit.

GPA is made as a separate unit

to ensure the best frequency stability Its frequency is tunable in the range of 7000 -7200 kHz. When receiving amateur radio stations in the range of 40 m, the first harmonic of the GPA signal is used, and in the range of 20 m - the second. When switching from range to range, only the input bandpass filters L1-L2-C2-C3-L3-C4 are switched.

Coils L1-L3 - ready-made, installed on the band bars (41 and 25 m) of the VEF-202 radio receiver. The number of turns is selected as follows. To the winding of the loop coil of the former local oscillator, the turns of the now unnecessary communication coil are wound (41 m range bar) and, conversely, the turns are unwound from the input coil on the 25 m range bar so that the “trimmers” of the coils can move freely, their thread must be moistened with alcohol.

The L4 GPA coil is wound on a ready-made factory frame 010 mm and 27 mm long. The frame has grooves for laying the wire. The number of turns is -12, the outlet is from the 4th turn. Wire - silver-plated 00.31-0.35 mm.

Setting up the receiver is reduced to the selection of parts indicated in the diagram with an asterisk, and laying the boundaries of the range of a smooth local oscillator. To adjust the band-pass filters, the knob of the capacitor C1 is displayed on the front panel of the receiver.

Of course, the receiver can be made multi-band - for example, using the former domestic broadcast radio receiver VEF-202 for this purpose with almost all of its own components (a vernier device with a variable capacitor, a drum switch with band strips, input and output connectors, and so on).

Schematic diagrams of a direct conversion receiver on transistors. Node assignment.

1. Radio frequency preselector amplifier.

The task of this block is to attenuate strong out-of-band interfering signals, side reception channels corresponding to the frequencies 2Fget., 3Fget. etc. and an increase in the minimum level of signals received in a given range to the level of the intrinsic noise of the converter (2), which contributes to an increase in the sensitivity of the receiver.

Preselector amplifier - circuit

Rice. 3. Scheme of a band-pass filter.

2. Frequency converter.

The converter directly transfers the radio frequency (RF) to the audio frequency (AF). It must have a high transmission coefficient, low noise level (to increase sensitivity). The design uses a mixer on counter-parallel diodes.

3. Local oscillator.

A local oscillator is a high-frequency oscillation generator of low power. The local oscillator largely determines the quality of radio reception. The first, very important requirement for the local oscillator is the high stability of its frequency. Any slight instability of the local oscillator will lead to a change in the tone of the telegraph or telephone signal spectrum. Another, no less important requirement is the absence of modulation of the local oscillator signal by noise, AC hum, and changes in the supply voltage. Smooth tuning of the local oscillator frequency is carried out using a variable capacitor.

The local oscillator circuit is shown in Fig. 4.

4. Low pass filter (LPF).

The low-pass filter must suppress low-frequency signals, the frequency of which is the upper limit of the speech spectrum (> 3 kHz). The quality of a filter is determined primarily by the number of filter units (order). The design of the receiver uses a single-section inductive-capacitive filter.

Low Pass Filter Diagram five.

5. Audio frequency amplifier (UZCH).

In a direct conversion receiver, almost all amplification occurs in the UHF. It should have a large gain, about 10 thousand. … 100 thousand. times, the lowest possible noise level, have enough power to ensure the operation of telephones or a loudspeaker. The ultrasonic frequency converter must be well protected from interference of electromagnetic waves directly to its input, interference from the power supply.

Audio frequency amplifier (UZCH). Rice. 6.

This design provides for receiving signals to headphones with a resistance of 50 ohms.

Construction and details.

List of denominations of used parts:

Preselector-amplifier, converter (1,2) see Fig.2.

Resistors (power 0.25 W):

  • R1 - 560 Ohm,
  • R2 - 10 Ohm,
  • R3 - 100 Ohm,
  • R4 - 10 Ohm,
  • R5 - 1.8 kOhm.

Capacitors:

  • C1 - 10 n,
  • C2 - 0.1 uF,
  • C3 - 10 n,
  • C4 - 10 n.

Diodes VD1, VD2 - KD503A.

Transistor VT1 - KT3102G.

  • Transformer T1 - on a ferrite ring 2000 NM, 18 turns of PEV-0.15, winding in three twisted wires.

Heterodyne. (3) Fig. 4.

Resistors:

  • R1 - 12 Kom,
  • R2 - 12 kOhm,
  • R3 - 680 Ohm,
  • R4 - 220 Ohm.

Capacitors:

  • C1 - 220 pF,
  • C2 - 5-50 pF KPI,
  • C3 - 220 pF,
  • C4 - 470 pF,
  • C5 - 510 pF,
  • C6 - 0.1 uF.

Diode VD1 - KS168A.

Transistor VT1 - KT315A.

Low pass filter (LPF). (4) fig. five.

Capacitors:

  • C1 - 47 n,
  • C2 - 47 n,

Choke T1 - on a ferrite ring 2000 NM, 250 turns PELSHO-0.12.

Audio frequency amplifier (UZCH) (5) fig.6.

Resistors:

  • R1 - potentiometer, 4.7 kOhm,
  • R2 - 22 kOhm,
  • R3 - 12 kOhm,
  • R4 - 10 kOhm,
  • R5 - 47 kOhm,
  • R6 - 47 kOhm,
  • R7 - 2.2 kOhm,
  • R8 - 12 kOhm,
  • R9 - 2.4 kOhm.

Capacitors:

  • C1 - 10 uF,
  • C2 - 4.7 uF,
  • C3 - 47 uF,
  • C4 - 10 uF.

Transistors:

  • VT1 - KT3102G,
  • VT2, VT3 - KT315A.

So, the radio receiver was tested at a collective radio station and showed good results: many Russian and foreign radio stations were heard. The receiver is great for a beginner radio amateur to observe the 40m band. Author of the work: Golubkin Nikolai Sergeevich, Rostov-on-Don.

Discuss the article DIRECT CONVERSION RECEIVER

Scheme of a simple HF observer receiver for any amateur radio band

Good day dear radio amateurs!
I welcome you to the site ""

Today we will consider a very simple, and at the same time providing good performance circuit - HF observer receiver - shortwave.
The scheme was developed by S. Andreev. I cannot help but note that no matter how much I have seen the developments of this author in amateur radio literature, they were all original, simple, with excellent characteristics and, most importantly, accessible for repetition by beginner radio amateurs.
A radio amateur's first step into the elements usually always begins with observing the work of other radio amateurs on the air. It is not enough to know the theory of amateur radio communication. Only by listening to amateur radio, delving into the basics and principles of radio communication, a radio amateur can gain practical skills in amateur radio communications. This scheme is just for those who want to take their first steps in amateur communications.

Presented radio amateur receiver circuit - shortwave very simple, made on the most affordable element base, easy to set up and at the same time providing good performance. Naturally, due to its simplicity, this circuit does not have “stunning” capabilities, but (for example, the sensitivity of the receiver is about 8 microvolts) will allow a novice radio amateur to comfortably study the principles of radio communication, especially in the 160 meter range:

The receiver, in principle, can operate in any amateur radio band - it all depends on the parameters of the input and heterodyne circuits. The author of this circuit tested the operation of the receiver only for the ranges of 160, 80 and 40 meters.
What range is better to assemble this receiver. To determine this, you need to take into account in which area you live and proceed from the characteristics of the amateur bands.
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The receiver is built according to the direct conversion scheme. It receives telegraph and telephone amateur stations - CW and SSB.

Antenna. The receiver works on an unmatched antenna in the form of a piece of mounting wire, which can be stretched diagonally under the ceiling of the room. For grounding, a pipe of the plumbing or heating system of the house, which is connected to terminal X4, is suitable. Antenna lowering is connected to terminal X1.

Principle of operation. The input signal is highlighted by the L1-C1 circuit, which is tuned to the middle of the received range. Then the signal goes to the mixer, made on 2 transistors VT1 and VT2, in a diode connection, connected in anti-parallel.
The voltage of the local oscillator, made on the transistor VT5, is supplied to the mixer through the capacitor C2. The local oscillator operates at a frequency two times lower than the frequency of the input signal. At the output of the mixer, at the connection point C2, a conversion product is formed - a signal of the difference between the input frequency and the doubled frequency of the local oscillator. Since the value of this signal should not be more than three kilohertz (the “human voice” fits in the range up to 3 kilohertz), then after the mixer the low-pass filter is turned on on the inductor L2 and capacitor C3, which suppresses the signal with a frequency above 3 kilohertz, due to which high selectivity of the receiver and the ability to receive CW and SSB. At the same time, AM and FM signals are practically not received, but this is not very important, because radio amateurs mainly use CW and SSB.
The selected low-frequency signal is fed to a two-stage low-frequency amplifier based on transistors VT3 and VT4, at the output of which high-resistance electromagnetic telephones of the TON-2 type are switched on. If you only have low-resistance phones, then they can be connected through a transitional transformer, for example, from a radio station. In addition, if a 1-2 kOhm resistor is connected in parallel with C7, then the signal from the VT4 collector through a capacitor with a capacity of 0.1-10 microfarads can be applied to the input of any ULF.
The supply voltage of the local oscillator is stabilized by the Zener diode VD1.

Details. You can use different variable capacitors in the receiver: 10-495, 5-240, 7-180 picofarads, it is desirable that they be with an air dielectric, but they will also work with a solid one.
For winding the loop coils (L1 and L3), frames with a diameter of 8 mm are used with threaded tuning cores made of carbonyl iron (frames from the IF circuits of old lamp or lamp-semiconductor TVs). The frames are disassembled, unwound and a cylindrical part 30 mm long is sawn off from them. Frames are installed in the holes of the board and fixed with epoxy glue. Coil L2 is wound on a ferrite ring with a diameter of 10-20 mm and contains 200 turns of PEV-0.12 wire wound in bulk, but evenly. The L2 coil can also be wound on the SB core and then placed inside the SB armor cups by gluing them with epoxy glue.
Schematic representation of mounting coils L1, L2 and L3 on the board:

Capacitors C1, C8, C9, C11, C12, C13 must be ceramic, tubular or disc.
Winding data of coils L1 and L3 (PEV wire 0.12) ratings of capacitors C1, C8 and C9 for different ranges and used variable capacitors:

The printed circuit board is made of foil fiberglass. The location of the printed tracks - on the one hand:

Establishment. The low-frequency amplifier of the receiver, with serviceable parts and error-free installation, does not need to be adjusted, since the operating modes of the transistors VT3 and VT4 are set automatically.
The main adjustment of the receiver is the establishment of a local oscillator.
First you need to check the presence of generation by the presence of RF voltage at the tap of the coil L3. Collector current VT5 must be within 1.5-3 mA (set by resistor R4). The presence of generation can be checked by changing this current when you touch the heterodyne circuit with your hands.
By adjusting the heterodyne circuit, it is necessary to provide the desired overlap of the local oscillator in frequency, the frequency of the local oscillator must be tuned within the ranges:
- 160 meters - 0.9-0.99 MHz
- 80 meters - 1.7-1.85 MHz
- 40 meters - 3.5-3.6 MHz
The easiest way to do this is to measure the frequency at the tap of coil L3 with a frequency meter capable of measuring frequencies up to 4 MHz. But you can also use a resonant wavemeter or an RF generator (beat method).
If you are using an RF generator, you can set up the input circuit at the same time. Apply a signal from the GHF to the receiver input (place the wire connected to X1 next to the generator output cable). The RF generator must be tuned within frequencies twice as high as indicated above (for example, in the range of 160 meters - 1.8-1.98 MHz), and the local oscillator circuit should be adjusted so that, with the corresponding position of the capacitor C10, a sound with a frequency of 0.5-1 kHz. Then, tune the generator to the middle of the range, tune the receiver to it, and adjust the L1-C1 circuit for the maximum sensitivity of the receiver. You can also calibrate the receiver scale using the generator.
In the absence of an RF generator, the input circuit can be set up to receive a signal from an amateur radio station operating as close to the middle of the range as possible.
In the process of tuning the circuits, it may be necessary to adjust the number of turns of the coils L1 and L3. capacitors C1, C9.

The direct conversion receiver for beginner radio amateurs is of unflagging interest. The design described works on the widely used bands 80m and 40m. Because there is a lot of interest in systems with direct frequency conversion. A direct conversion receiver circuit was developed for 80 and 40 meters. On inexpensive and popular parts that can be found in almost every radio amateur in the box. With good transmission, the receiver provides reception of both telegraph (CW) and (SSB) signals in the 3.5-17 MHz bands. One of the disadvantages resulting from direct conversion is the reception of two signals.

How it works?

The principle of direct frequency conversion has already been explained many times. But it should be remembered that the acoustic signal is obtained as the difference between the frequencies of the input signal and the signal from the generator.

The direct conversion receiver circuit diagram is shown in the figure.

– range switch PZ1 80 / 40m

Switching of LC (input and generator) circuits

– potentiometers: P1 (volume), P2 (coarse), P3 (fine)

– T5 transistor for low-noise headphone connection

– PZ2 power switch with Li-Ion 2×3.7V batteries (allows switching from an external 12V power supply to an internal power supply)

After that, we will follow the signal path in the circuit with direct conversion from the antenna to the headphones. Input P1 has the function of an attenuator and at the same time a volume control at the antenna input. The next element is the resonant circuit, this is an input filter of 40 m, filtering the signal from the antenna to the input of the amplifier transistor T1 (switch PZ1 in the upper position, as in the diagram). Capacitor C1, together with the main coil L1, creates a resonant circuit at a frequency of about 7.1 MHz. After setting switch PZ1 to the down position, capacitor C1 will be connected to capacitor C17, changing the frequency of the resonant circuit by about 3.7 MHz.

The input signal after amplification with T1 is directed to a mixer consisting of two pulse diodes D1-D2 connected in opposite parallel. The system works like a key, closing the circuit at twice the frequency of the generator. An important property of such a mixer is that the oscillator must be tuned to twice the frequency of the input signal, which is very important because of the greater stability of the oscillator and the less ability of the oscillator signal to penetrate the antenna.

Potentiometer R4 is used to accurately balance the detector. The YFO on transistor T2 supplies the detector with a signal in the range of 3500-3600 kHz for the 40 m range and 1750-1900 kHz for the 80 m range.

The operating frequency of the generator is determined by the operating frequency of the L2C5 circuit. Coil L2 is tapped from the middle of the winding and operates in the range of 40 m (the lower half is closed to ground using the second section of the PZ1 switch, as in the diagram). The oscillator frequency setting is implemented using a D3 varicap type BB112.

In this case, tuning occurs by changing the voltage applied to the varicap cathode from the potentiometer P2 (basic setting). The additional potentiometer P3 functions as a simple precision tuner. Which provides fine tuning of the received station (the tuning range is not constant and is the largest in the upper part of the frequency range). The best solution for tuning accuracy and comfort would be to use a multi-turn potentiometer, but - without using a dial - you can't even determine the approximate receiving frequency.

Frequency calibration from above (7.2 MHz reception) allows the use of capacitor C19. An additional capacitor C18 is useful when calibrating the 3.8 MHz frequency (may not be necessary if the number of coil turns is chosen accurately).

The generator setting range in the 40 m band is limited from below by resistor R16.

After setting the switch PZ1 to the lower position (range 80 m), the entire winding L2 works, and the setting range is increased by adding an additional resistor R14. With correctly set generator ranges in the extreme positions of potentiometer P2, reception of amateur bands of 3.5-3.8 MHz and 7.0-7.2 MHz is obtained.

On the next two transistors T3 and T4, a two-stage bass amplifier is built. To connect headphones at the output, an additional emitter follower on a T5 transistor was added. When using stereo headphones, connect them in parallel through the appropriate pin connection in the headphone jack.

The way the receiver is powered, thanks to the PZ2 switch, it is possible to power it from an external power supply of about 12 V or from internal batteries, which is convenient. For example, when working in the field or eliminating interference from the mains power supply.

In either case, the oscillator circuit is powered by a stabilized 5V output from the 78L05 regulated power supply.

Assembly and commissioning of a direct conversion receiver for 80 and 40 meters.

The entire receiver circuit was assembled on a single-sided board (figure).

Of course, such a board can be prepared manually, take foil fiberglass with dimensions of 100 × 75 mm, cut out in the form of squares with sides of about 8 mm. Such platforms, isolated from a common wire, can be made in any way (etching, milling or cutting).

The assembly of the receiver elements on the printed circuit board is shown in the figure.

The other side of the board has the internal power supply and all the controls and connectors.

The connectors (antenna, power and headphones) were attached to the back of the receiver, and the potentiometers (P1, P2, P3) were mounted on the front panel. On the left, a PZ1 range switch was installed next to the coils L1 and L2. The body of the receiver was made of fiberglass strips 40 mm high, soldered together with the circuit board. The upper and lower parts of the body can also be made of fiberglass or aluminum sheet. Of course, everyone can choose a different metal case, but the proposed design serves its purpose well.

In any case, it is advisable to assemble the elements after preparing all the body components and attaching the adjusting elements and sockets. The coils of the circuit are the most complex, so you should pay special attention to them, since the parameters of the receiver depend mainly on them.

The receiver coils were wound with 0.4 wire on two T50-2 toroidal cores with an outer diameter of 12.7 mm. These are red cores with dimensions of 12.7×7.7×4.83 mm and AL = 4.9. Antenna coil L1 (5uH) contains 32 turns with a tap from 6 turns from the common wire connection, and a coupling coil L1 (same wire). The generator coil L2 (12.5uH) contains 50 turns of wire with a tap in the middle, that is, after the 25th coil turn (about 3.2uH). All windings should be evenly distributed around the entire circumference, and after winding, it is recommended to check them with an inductance meter or multimeter.

When powering up the circuit, first check the collector voltages of the transistors if they are close to about half the supply voltage. In case of significant differences (which may occur with the use of transistors and other gain), the base resistors must be adjusted.

After making sure that the operating voltages of all transistors are set correctly, it is necessary to check the generator. The output frequency of the receiver can be checked with a frequency meter connected through a capacitor of about 20 pF, for example, with resistor R4 or an additional receiver (with a short wire antenna) similar to our receiver (resistor R6 must be set to maximum signal at the beginning). To obtain the lower and upper ranges, perform the following operations at the extreme positions of the main tuning knob.

First set slider P2 to the far right (P3 could be in the middle) and switch PZ1 to 80m. If the diode cathode voltage is close to 5V, the ends of potentiometer P2 should be replaced.

With these settings, the oscillator frequency should be slightly higher than 1.9 MHz. If the frequency does not match, we correct it with a capacitor (C19) exactly to the value of 1900 kHz, which corresponds to the received frequency of 3.8 MHz. If this cannot be achieved with a capacitor, you will need to adjust the C5 capacitor (reducing will increase the frequency). If there is a desire to correct the number of turns of the L2 coil, this must be done symmetrically, that is, on both sides of the tap.

After moving PZ1 up to 40 m, the frequency should be close to 3.6 MHz. It is better if it is a little higher, because then it can be easily adjusted by selecting the capacitor C18. It may also be necessary to move the tap, which is physically not so easy, because then you have to wind on one side, and on the other - unwind the same part of the turns of the coil. In any case, you need to get exactly 3600 kHz, which corresponds to the received frequency of 7.2 MHz. It may happen that the previously set value of 1900 kHz has changed, so you need to fix it again until it works.

Setting lower frequencies will be easier if you first turn on the setting potentiometer R16 instead of eg 47k. After setting P2 to the leftmost position and PZ1 to 40 m, the value of R16 should be chosen so that the oscillator frequency is 3500 kHz, which corresponds to the received frequency of 7.0 MHz. In turn, after moving PZ1 to 80 m, the value of R14 must be chosen so that the generator frequency is 1750 kHz (the resulting frequency is 3.5 MHz).

If it is not possible to get the tuning of the lower ranges in the way that telegraph stations work, this means that the tuning range is too small, then the capacitor C20 must be increased, but the whole tuning operation must be carried out again. This tried and tested procedure will also be useful when tuning narrower bands limited to, for example, the most used site SSB. In this case, instead of the D3 BB112 varicap, you can use another option with a smaller range (two BB105 diodes may be enough).

With the oscillator tuned, the final step in setting up the receiver is to check it works with the antenna connected. It is also worth trying to select the value of capacitor C1 for the strongest received station signal in the middle of the 40m range. Finally, set the R6 slider to the best signal to noise ratio.

The last step is to create a frequency timeline around the P2 potentiometer

Potentiometer R4, used to accurately balance the detector, you can set the minimum signal on the resistor R3 using, for example, an RF probe to a multimeter.

When pairing, the diodes R4 can be omitted, for example, by shorting parts of the wire. The receiver with a dual-band 80/40 m antenna made it possible to receive a sufficient number of local and foreign CW / SSB stations. Antenna dipole: 2×19.5 m, connected by one coaxial cable.

It so happened that at certain times and under special conditions of radio wave propagation, it was possible to hear stations in the receiver at a frequency of 40 m, regardless of the frequency setting. This undesirable effect is reduced by turning on the P1 attenuator. The use of this attenuator was also necessary in the case of a close, strong neighbor radio station. The amateur bands 80m and 40m during the day are generally suitable for short range radio communications. At night, these bands "open" and you can listen to European countries and even stations from other continents (DX).